Method and apparatus for controlling a switched reluctance machine

ABSTRACT

A method and apparatus for controlling the average voltage applied to a phase winding of a reluctance machine during a period of current increase, controlling the current when a desired current is reached, and controlling the voltage applied during a period of current decrease, to improve machine efficiency and thereby reduce acoustical noise produced by the machine, and unwanted vibration. The apparatus comprises circuitry and associated elements which are operable to switch the phase winding into circuits which apply either a positive DC voltage, a zero voltage, or a negative DC voltage, to increase, maintain, or decrease the current in the winding. The timing and the rate of the switching is controlled as a function of the angular position and speed of a rotor of the reluctance machine, the torque output of the machine, and the desired speed or torque. 
     A method is provided whereby a duty cycle corresponding to a specific reluctance machine, as well as to specific operating parameters of the machine at a given rotor position, speed, and torque, is calculated by the controller for applying voltage to the phase winding during the period of current increase to increase the current from at or near zero to a desired peak current. Additionally, a method is provided for calculating a duty cycle for the time at which the phase is de-energized, corresponding to a specific reluctance machine, and to specific operating parameters of the machine at a given rotor position, speed, and torque, as well as a method for changing the duty cycle to drive the current in the winding to zero or near zero. The calculation of the appropriate duty cycle for the specific operating conditions of the reluctance machine controls the net average voltage applied, has been found to reduce abrupt changes in the current. This reduction correlates into reduced abrupt changes in the magnetic flux, resulting in reduced acoustic noise and unwanted vibration.

This is a divisional of co-pending Ser. No. 08/635,240 filed Apr. 12,1996.

FIELD OF THE INVENTION

The present invention relates to machine systems and more particularlyto a method and apparatus for controlling the voltage and currents inthe phase windings of a machine, such as a switched reluctance machine,to improve machine performance by, for example, reducing unwanted noiseand vibration.

BACKGROUND OF THE INVENTION

In general, a reluctance machine is an electric machine in which torqueis produced by the tendency of its movable part to move into a positionwhere the reluctance of an excited winding is minimized (i.e., theinductance is maximized).

In one type of reluctance machine the energization of the phase windingsoccurs at a controlled frequency. These machines are generally referredto as synchronous reluctance machines. In a second type of reluctancemachine, circuitry is provided for detecting the angular position of therotor and energizing the phase windings as a function of the rotor'sposition. This second type of reluctance machine is generally known as aswitched reluctance machine. Although the description of the presentinvention is in the context of a switched reluctance machine, thepresent invention is applicable to all forms of reluctance machines,including synchronous and switched reluctance motors, synchronous andswitched reluctance generators, as well as to other machines that havephase winding arrangements similar to those of switched reluctancemachines.

The general theory of design and operation of switched reluctancemachines is well known and discussed, for example in TheCharacteristics, Design and Applications of Switched Reluctance Motorsand Drives, by Stephenson and Blake and presented at the PCIM '93Conference and Exhibition at Nuremberg, Germany, Jun. 21-24, 1993.

When a switched reluctance machine is running, including at low speedsor a standstill, the torque (and other machine performance parameters)may be adjusted by monitoring the rotor's position, energizing one ormore phase windings when the rotor is at a first angular position,referred to as the "turn-on angle (T_(ON),)" and then de-energizing theenergized windings when the rotor rotates to a second angular position,referred to as the "turn-off angle (T_(OFF))." The angular distancebetween the turn-on angle and the turn-off angle is often referred to asthe "conduction angle."

At standstill and at low speeds, the torque of a switched reluctancemachine can be controlled by varying the magnitude of the current in theenergized phase windings over the period defined by T_(ON) and T_(OFF).Such current control can be achieved by chopping the current using acurrent reference with phase current feedback. Such current control isreferred to as "chopping mode" current control. Alternately, pulse widthmodulation (PWM) voltage control may be used.

As the angular speed of the motor increases, a point is reached wherethe amount of current which can be delivered into a phase winding duringeach phase period is limited by the rapidly increasing inductance andcounter emf associated with the winding. At such speeds pulse widthmodulation or chopping strategies are less desirable and the torque ofthe machine is commonly controlled by controlling the duration of thevoltage pulse applied to the winding during the phase period withrespect to the rotor's position. Because a single pulse of voltage isapplied during each phase period, this form of control is often referredto as "single-pulse control."

As a switched reluctance motor (or generator) operates, magnetic flux iscontinuously increasing and decreasing in different parts of themachine. This changing flux will occur in both chopping mode andsingle-pulse current control. The changing flux results in fluctuatingmagnetic forces being applied to the ferromagnetic parts of the machine.These forces can produce unwanted noise and vibration. One majormechanism by which these forces can create noise is the ovalizing of thestator caused by magnetic forces normal to the air-gap. Generally, asthe magnetic flux increases along a given diameter of the stator, thestator is pulled into an oval shape by the magnetic forces. As themagnetic flux decreases, the stator pulls or springs back to itsundistorted shape. This ovalizing and springing back of the stator willproduce audible noise and can cause unwanted vibration.

In addition to the distortions of the stator by the ovalizing magneticforces, acoustic noise and unwanted vibration may also be produced byabrupt changes in the magnetic forces in the motor. These abrupt changesin the gradient of the magnetic flux (i.e., the rate of change of theflux with time) are referred to as "hammer blows" because the effect onthe stator is similar to that of a hammer strike. Just as a hammerstrike may cause the stator to vibrate at one or more naturalfrequencies (determined by the mass and elasticity of the stator) theabrupt application or removal of magnetic force can cause the stator tovibrate at one or more of its natural frequencies. In general, thelowest (or fundamental) natural frequency dominates the vibration,although higher harmonics may be emphasized by repeated excitation atthe appropriate frequency.

In addition to the stator distortions resulting from the ovalizing andhammer blow phenomena described above, the fluctuating magnetic forcesin the motor can distort the stator in other ways, as well as distortingthe rotor and other parts of the machine system. For example,distortions of the rotor can cause resonance of the rotor end-shields.These additional distortions are another potential source of unwantedvibration and noise.

Although the problem of unwanted acoustic noise and vibration has beenrecognized, known control systems for reluctance motors do notadequately solve the problem. For example, the general problem ofacoustic noise in switched reluctance motor systems is discussed in C.Y. Wu and C. Pollock, "Analysis and Reduction of Vibration and AcousticNoise in the Switched Reluctance Drive," Proceedings of the IAS '93 pp.106-113 (1993). In general, the method suggested by Wu and Pollockinvolves control of the current in the phase winding such that thecurrent is controlled in two successive switching steps with the secondswitching step occurring approximately one-half of a resonant cycleafter the first where the resonant cycle is defined by the naturalfrequency of the machine. This approach is typically implemented byswitching off one of the power devices at a first point in time to causea first stepped reduction in applied voltage, and then later switchingoff the second power device. Between the time when the first switchingdevice is switched off and the second switching device is switched off,the current is allowed to freewheel through a freewheeling diode and thesecond switching device.

The two-step voltage-reduction approach to noise reduction in switchedreluctance motors discussed above suffers from several limitations anddisadvantages. One such limitation is that in many cases the two-stepvoltage-reduction approach requires precise switching of the switchingdevices within the interval defined by the turn-on and turn-off angles(i.e., the angular interval during which the phase winding isenergized). Still further, the two-step voltage-reduction approachlimits the flexibility to dynamically adjust the freewheeling period foreach phase cycle. As discussed above, in the two-step voltage-reductionapproach, the duration of the freewheeling period is optimized to reducethe noise produced by the system at a single fundamental frequency.There are many instances when it would be desirable to optimize thefreewheeling duration according to other criteria.

An additional limitation of the two-step voltage-reduction approach, andother approaches that utilize freewheeling to reduce noise, is that,since there is typically only one freewheeling period per phaseenergization cycle, freewheeling generally reduces noise produced byonly a single frequency of the motor system. Freewheeling to reducenoise at one frequency does not reduce noise produced at otherfrequencies, in motor systems that have multiple resonant frequencies.Accordingly, such approaches do not reduce many sources of unwantednoise. A further disadvantage with the freewheeling approaches is thatthere are several motor control systems (e.g., H-circuits with a splitcapacitor, third-rail circuits, ring circuits and the like) that simplydo not allow freewheeling. These systems cannot use freewheeling toreduce noise.

The present invention overcomes many of the limitations anddisadvantages associated with known systems and provides a unique methodand apparatus for controlling the phase voltage and the phase windingcurrents in a phase windings of a switched reluctance machine to, forexample, reduce unwanted machine noise and vibrations.

SUMMARY OF THE INVENTION

One aspect of the present invention concerns a method and apparatus forcontrolling the average voltage applied to a phase winding of a machine,particularly a reluctance machine, during a period of current increase,and a period of current decrease, to improve machine performance byreducing abrupt changes in the phase current and the magnetic fluxassociated with the phase or with one or more adjacent energized phases.One aspect of this improved performance is a reduction in audible noiseand vibration generated by the machine. Audible noise and vibration aregenerated in a reluctance machine in part as a result of rapid changesin the magnetic flux which tend to distort or ovalize machine parts.Controlling the rate of change in the voltage and current in phasewindings during the periods when a winding is energized from a zero ornear-zero voltage state to a maximum voltage state, or from a maximumvoltage state to a zero or near-zero voltage state, can result in lessrapid change in the magnetic flux, thereby reducing audible noise andvibration.

An exemplary apparatus of the present invention concerns a controllerand associated circuitry. The method of the present invention concerns aseries of steps performed by the controller which govern the duty cycleof voltage pulses applied to a phase winding during current increase andcurrent decrease, as a function of the speed and torque output andrequirements of the reluctance machine.

The controller contains circuitry which can operate in several distinctmodes. The different modes may be selected based on the outputrequirements of the system, e.g. speed or torque requirements. Thecontroller may operate in a single mode, or may be programmed to changemodes based on the actual operating characteristics of the machine. Thedifferent modes define the optimum rotor angular positions at which thephase windings are energized and de-energized, in accordance with thedesired speed and/or torque performance of the machine. The controlleralso contains circuitry which controls the voltage applied to the phaseat the time the windings are energized, until a desired maximum currentis reached. The circuitry may include a microprocessor which determinesthe maximum current desired in the phase winding based on feedbacksignals representing the actual speed and torque of the machine.

In one exemplary embodiment, from the time when the phase is energized,until the time when the maximum current is reached in the phase, theaverage voltage applied to the winding is controlled using soft-choppingtechniques, i.e., switching the phase winding between a circuit whichapplies a positive DC voltage and a circuit which applies a zerovoltage, thereby allowing the current in the phase winding to freewheeland decrease slowly. In this embodiment, the soft-chopping duty cycle iscalculated as a function of the angular speed of the rotor and thetorque generated by the machine, and the desired speed or torque of themachine. As a result, the gradient of the increasing current in thephase winding will vary with the speed and torque of the machine.

In another embodiment, from the time when the phase is at its desiredmaximum current, until the time when the phase is de-energized,soft-chopping techniques are continued to maintain the current at thedesired level. Alternative methods of controlling the current level canbe applied, such as hard-chopping techniques, whereby the phase isswitched between a circuit which applies a positive DC voltage and acircuit which applies a negative DC voltage, thereby quickly driving thecurrent in the winding to zero.

In an exemplary embodiment, at the time when the phase is de-energized,until the time when the current in the phase is at or near zero, theaverage voltage applied to the winding is controlled by usinghard-chopping techniques. An aspect of this embodiment is to calculatean initial duty cycle for the applied voltage pulses which is a functionof the last soft-chopping duty cycle, and to apply a net average voltageto the phase winding in the first hard-chopping duty cycle which is thesame as the net average voltage at the time the phase was de-energized.It is also an aspect of this embodiment that the duty cycle of thehard-chopping pulses, during the time period when the current in thephase is decreasing, is modified and controlled such that the netaverage voltage applied to the winding changes from a positive DCvoltage, to zero voltage, to a negative voltage, until the current inthe windings reaches zero or a desired value near zero. The rate ofchange of the duty cycle varies as a function of the angular speed ofthe rotor and the torque generated by the machine, and the desired speedor torque of the machine. As a result, the gradient of the decreasingcurrent in the phase winding will vary with the speed and torque of themachine.

In other exemplary embodiments, where the current in the phase prior tode-energization is controlled by either hard chopping or soft chopping,the last complete duty cycle or net average voltage prior tode-energization is captured and the same hard chopping or soft choppingswitching strategy is continued with the duty cycle ramped down from apositive net average voltage to zero or a net negative voltage.Thereafter, at one or more predetermined times or at a one or morepreselected rotor positions during the period where the duty cycle isramped down, the duty cycle may be captured again, and the switchingstrategy may be changed again. This could include changing switchingstrategies such that a hard chopping duty cycle is applied, startingwith the same net average voltage as the prior duty cycle but rampeddown at a different rate, or actuating both switches into anon-conducting state thereby applying the full negative voltage. It willbe evident to one skilled in the art that the controller of the presentinvention can combine the various possible switching strategies duringthe period after the phase winding is de-energized in many possiblecombinations other than those discussed above.

The controller of the invention can perform these techniques forcontrolling the rate of current increase and current decrease in asingle phase winding of a reluctance machine or other device in whichcurrent is controlled as a function of a rotor position, or in multiplephase windings. The control apparatus and methods are applicable to anymethod of operation, and any method of determining the rotor positionsat which the phase is energized and de-energized, since the rotor speedand machine torque are factored into the duty cycle of the appliedvoltage pulses required.

BRIEF DESCRIPTION OF THE DRAWINGS

Other aspects and advantages of the present invention will becomeapparent upon reading the following detailed description and uponreference to the drawings in which:

FIG. 1 illustrates a reluctance machine system in accordance with thepresent invention.

FIG. 2 illustrates in greater detail the reluctance machine 20 of FIG.1.

FIG. 3A illustrates in greater detail the power converter 30 of FIG. 1.

FIG. 3B illustrates the types of voltages that may be established acrossa phase winding in a reluctance machine by power connector 30.

FIGS. 4A and 4B illustrate in greater detail the controller 40 of FIG.1.

FIG. 5A illustrates in flowchart form a method used by controller 40 forselecting a hard-chopping duty cycle for chopping at the T_(OFF) rotorposition.

FIG. 5B illustrates a typical maximum applied average voltage (definedby a simplified HARD-RAMP-START duty cycle) vs. torque outputcalculation for various speeds.

FIG. 5C illustrates voltage applied to a phase winding by the method andapparatus of the current invention and the resulting current waveformgenerated in the phase.

FIG. 5D illustrates the net average voltage applied to a phase windingby the method and apparatus of the current invention and the resultingcurrent waveform generated in the phase.

FIG. 6A illustrates in flowchart form one method in which controller 40performs a controlled RAMP-DOWN of the phase current by graduallyreducing the net average voltage applied to the phase wing.

FIG. 6B illustrates exemplary curves for selecting a RAMP-DOWN-GRADIENT,which defines the rate of change in the net average voltage applied tothe winding, as a function of machine speed and torque.

FIG. 6C illustrates exemplary circuitry for implementing the RAMP-DOWNmethod of the present invention.

FIG. 7A illustrates an alternate embodiment of the invention wherein a+HVDC to zero voltage soft-chopping switching strategy is employed atT_(OFF), the duty cycle is ramped down to zero, and then a hard-choppingswitching strategy is employed.

FIG. 7B illustrates another alternate embodiment of the inventionwherein a +HVDC to zero voltage soft-chopping switching strategy isemployed at T_(OFF), the duty cycle is ramped down to zero, and then a-HVDC to zero voltage soft-chopping switching strategy is employed.

FIG. 8A illustrates in flowchart form a method in which controller 40performs (1) a controlled RAMP-UP of the phase current and (ii)efficient fixed frequency chopping circuit in the active cycle.

FIG. 8B illustrates current in a phase winding and the duty cycle of theapplied voltage pulses during RAMP-UP.

FIG. 8C illustrates exemplary circuitry for implementing the RAMP-UP andRAMP-DOWN control methods of the present invention.

FIGS. 9A and 9B compare a traditional current waveform (FIG. 9A) with acurrent waveform generated through use of the methods and apparatus ofthe present invention (FIG. 9B).

Similar reference characters indicate similar parts throughout theseveral views of the drawings.

DETAILED DESCRIPTION OF THE INVENTION

Although the present invention is discussed in terms of a system whichincludes a reluctance machine, the method and apparatus may be appliedto many other machine systems in which current is applied to a windingor coil as a function of a rotor or armature position, including variousinductance motor systems, stepper motor systems, and other motor orgenerator systems. Turning to the drawings and referring to FIG. 1, oneexample of a system 10 that may be used to implement the methods of thepresent invention is illustrated in block form.

As illustrated, an electronic controller 40 receives signals from arotor position transducer ("RPT") 3. The RPT 3 detects the angularposition of the rotor of a reluctance machine 20 and provides acontroller 40 signals representative of the rotor's position. RPT 3 maycomprise optical or magnetic sensor(s) and may be of conventionalconrtruction. Embodiments are envisioned wherein the RPT is replaced bycircuitry that monitors the electrical characteristics of the phasewinding and provides signals representative of the rotor's angularposition and/or speed without the use of position sensors. One suchsensorless approach is disclosed in the currently pending applicationSerial No. 08/549457, "Rotor Position Sensing in a DynamoelectricMachine Using Coupling Between Machine Coils" filed Nov. 27, 1995,assigned to the assignee of the present invention.

In the embodiment of FIG. 1 the controller 40 derives an indication ofthe angular speed of the rotor of the reluctance machine 20 byappropriately processing the position information provided by the RPT 3.Alternate embodiments are envisioned wherein a separate tachometer orsimilar device provides speed information to the controller 40.

In addition to receiving signals from RPT 3 concerning the rotor'sposition and angular speed of the machine, controller 40 also receivesvia line 4 feedback signals from a power converter 30. In oneembodiment, the feedback signals represent the currents that acesupplied by the power converter 30 to the phase windings of the rotor.The controller also receives an externally generated signal on line 2corresponding to the required torque or speed of the machine 20.

In response to the rotor position signals from RPT 3, the feedbacksignals from a power converter 30 and the torque/speed command on line2, the controller 40 provides switching signals (sometimes referred toas "firing signals") via lines 4a-4c to a number of power switchingdevices that comprise a power converter 30. The switching devices in thepower converter 30 are connected via lines 5a-5c to three phase windingsA, B and C of a three phase switched reluctance machine 20. The threephase windings of the machine (A, B, and C) are schematicallyillustrated in FIG. 1. Those skilled in the art will recognize thatalthough a three phase machine has been shown for illustrative purposes,machines with more or less than three phase windings can be used. Thepresent invention applies equally to such machines.

In general, the electronic controller 40 responds to the positionsignals from the RPT 3 by generating firing signals for each of thethree phases of the motor to switch the power devices in power converter30 such that the phase windings A, B and C are energized in the propersequence over angular periods of rotor rotation to produce torque in adesired direction.

FIG. 2 illustrates in greater detail the three phase reluctance machine20 of FIG. 1.

In general the reluctance machine 20 consists of a stator 22 that isconstructed according to known techniques from a stack of statorlaminations that define twelve inwardly projecting stator poles 23.These poles define a principal stator axis (extending outwardly fromFIG. 2). A rotor 24 is coupled to a rotatable shaft (not shown) that isco-axial with the principal axis of the stator. The shaft is mounted onbearings and is free to rotate. The rotor 24 is formed from a stack ofrotor laminations that define eight outwardly projecting rotor poles 25.

Associated with each stator pole 23 is a wound coil of wire 26a, 26b and26c. The coils of opposing stator poles are placed such that currentflowing in the opposing stator poles at the same time will establishinwardly pointing electro-magnetics of opposite polarities.

In the reluctance machine of FIG. 2, sets of coils associated withopposing stator poles 23 are coupled together to form three phasewindings A, B and C where each phase winding is comprised of four coils26 and where each phase is associated with four stator poles 23. Theterminating ends of the three phase windings A, B and C are coupled tothe output of power converter 30 via connection lines 5a, 5b and 5c.

When electric current flows though the phase windings of a given phase(e.g., phase winding A) two sets of opposing electro-magnetics ofopposite polarity will be established in the machine. This isillustrated in FIG. 2 by the references to Phase A_(NORTH) and PhaseA_(SOUTH) electro-magnets which will be established when electriccurrent is flowing in a given direction in the phase winding A. Theelectro-magnets established by this current will produce a force ofattraction between the energized stator poles 23 and the rotor poles 25which will produce a torque. By switching energization from one phasewinding to another, the desired torque may be maintained regardless ofthe angular position of the rotor. By switching the energization of thephase windings to develop positive torque, the machine may be operatedas a motor; by energization of the phase windings to develop a negativetorque the machine may be operated as a brake or generator. Themagnitude of the produced torque can be controlled by controlling thecurrent in the energized phase winding which controls the strength ofthe established electro-magnetic field.

For the sake of illustration, a three phase machine having twelve statorpoles and eight rotor poles (i.e. a 12/8 machine) has been shown. Thoseskilled in the art will recognize that other combinations (e.g., 6/2,6/4, etc.) are possible and that machines with more or less than threephase windings can be used. The present invention applies equally tosuch machines. Moreover, the present invention is applicable to invertedmachines, where the stator is positioned within the bore of an outerrotating rotor, and to linear machines, where the rotor moves linearlywith respect to the stator.

As discussed in connection with FIG. 1, the phase windings of thereluctance machine are energized by the application of power to thephase windings by power converter 30. Power converter 30 is illustratedin greater detail in FIG. 3A.

Turing to FIG. 3A, AC power is supplied to the power converter 30 byinput power lines AC+ and AC-. An earth ground may also be applied tothe power converter 30 via a GND line. An appropriately sizedelectro-magnetic interference ("EMI") filter 31 receives and conditionsthe input AC power before supplying it to a full-wave rectifier 32.

Although not illustrated in FIG. 3A, front-end power protectioncircuitry may be used to prevent damage to the components of the powerconverter 30. For example, a fuse rated to the maximum allowableamperage may be placed in series with one of the AC+ or AC-power linesor a motor thermal cut off may be coupled to the EMI filter 31 to cutpower off to the reluctance machine 20 in the event that excessivelyhigh temperatures are detected in the vicinity of the machine 20 or thepower converter 30.

The full-wave rectifier 32 receives the AC power and converts in to DCpower such that high voltage DC power is available on high voltage DCbus lines +HVDC and HVDC_(Common). Parallel connected DC bus capacitors34 are used to filter the output of the fill wave rectifier 32 and toprovide a DC voltage across the high voltage DC bus. While the absolutemagnitude of the DC voltage provided across the DC bus will varydepending of the power rating of a given power converter the DC voltageacross the high voltage DC bus can reach levels of 160 Volts for 110Volt AC input and over 300 Volts for 220 Volt AC input.

Auxiliary DC power supplies consisting of a string of appropriatelysized resistors may be driven from the high voltage DC bus. In theexample of FIG. 3A a 5 Volt Vcc signal for powering the controller 40 isprovided by resistor chain 33a. A similar resistor chain 33b provides asource of 5 Volt power to the RPT 3.

Although not illustrated in FIG. 3A additional EMI or appropriate undervoltage detectors (e.g., a resistive chain with a resistive/capacitiveload) could be coupled across the high voltage DC bus. Such elements arenot particularly related to the subject matter of this disclosure andare within the understanding of one of ordinary skill in the art and arethus not discussed in detail herein.

Coupled across the high voltage DC bus are power switching devices anddiode "groups" where each group is associated with one of the threephases of the reluctance machine 20. Specifically, each group includesan upper power switching device 35, a lower power switching device 36,an upper fly-back diode 37 and a lower fly-back diode 38. A currentmeasuring sense resistor arrangement 39 is associated with each of thepower switching device groups. These current measuring sense resistorarrangements 39a, 39b and 39c provide voltages that corresponds to thecurrent flowing through the phase windings associated with eacharrangement. As discussed more fully below, these "current feedback"signals may be used by the controller 40 to control the current in thephase windings.

In the embodiment of FIG. 3A, each of the current measuring senseresistor arrangements 39a, 39b and 39c consists of a number of parallelconnected resistors. This arrangement is beneficial in some applicationsbecause it allows for the use of lower power rated (and thus lessexpensive) resistors. A single resistor of appropriate value could alsobe used.

The power switching devices 35 and 36 may comprise any suitable powerswitching devices such as MOSFETs, IGBTs, bi-polar transistors, SCRs ora combination of the above. For example, if only the upper switches willbe modulated for current control, then fast switching power MOSFETs canbe used for the upper switching devices 35 and slower switching (butlower loss) IGBT devices can be used for the lower power switchingdevices 36.

By controlling the actuation of the power switching devices 35 or 36,different voltage potentials can be established across the phase windingto which the appropriate group is coupled.

FIG. 3B illustrates the types of voltages that may be established acrossa phase winding coupled to one of the power switching groups of FIG. 3A.Using the Phase A group as an example, FIG. 3B(I) illustrates thevoltage that will be established across Phase winding A when both theupper power switching device 35a and the lower power switching device36a are actuated into a conducting (or "closed") condition. Whenswitches 35a and 36a are so actuated, the phase winding is coupledacross the high voltage DC bus and (ignoring losses introduced by theswitching devices) the voltage across Phase A winding is substantiallyequal to the high voltage DC bus voltage (+HVDC).

FIG. 3B(II) illustrates the voltage that will appear across the Phase Awinding when both of the power switching devices 35a and 36a areactuated into a non-conducting (or "open") configuration when current isflowing in Phase winding A. Because of the large inductance of the Phasewinding, the current in the winding cannot instantaneously change andthe current will thus continue to flow through the winding via thefly-back diodes 37a and 38a. Because both the upper and lower fly-backdiodes are conducting in this arrangement, the voltage applied acrossPhase winding A winding is substantially the negative of the highvoltage DC bus voltage (-HVDC).

FIG. 3B(III) illustrates the voltage that will appear across Phasewinding A when either the upper or the lower power switching devices 35aor 36a (but not both) is actuated into a non-conducting state while theother is in a conducting state, and current is flowing in the phasewinding. In this arrangement, because the current cannot instantaneouslychange, the current will continue to flow through the conducting powerswitching device and through the fly-back diode associated with thatpower conducting device. For example, if the lower power switchingdevice 36a is actuated to a non-conductive state, current will continueto flow through the upper power switching device 35a and through theupper fly-back diode 37a. Similarly if the upper switching device 35a isactuated to a non-conductive state, current will continue to flowthrough the lower power switching device 36a and lower fly-back diode38a. In either arrangement, the total voltage across the Phase A winding(again ignoring losses introduced by the switching devices and diodes)will be approximately zero volts. This arrangement, where current iscirculating through the winding with approximately zero volts across thewinding, is sometimes referred to as a "freewheeling" arrangement sincethe current "freewheels" through the winding.

By controlling the actuation of the power switching devices it ispossible to implement various switching schemes. For example, asillustrated in FIG. 3B(IV), the upper and lower switching devices canboth be alternately actuated to be conductive and non-conductive,simultaneously causing the voltage across the phase winding to vary from+HVDC to -HVDC. This scheme is sometimes referred to as "hard-chopping."By appropriately selecting the ratio of time for which the +HVDC voltageis applied to the phase winding as opposed to the -HVDC voltage, the netaverage voltage applied to the phase winding may be either positive ornegative such that the current in the phase winding may be controlled toincrease or decrease.

FIG. 3B(V) illustrates an alternate switching scheme that may be used tocontrol the average voltage applied to the phase winding. In thisswitching scheme, either: (i) the upper power switching device 35a isleft in a conducting state while the lower power switching device 36a isalternately switched from a conducting to a non-conducting state or (ii)the lower power switching device is left in a conducting state while theupper switching device 35a is alternately actuated between conductiveand non-conductive states. In addition, the power switching device thatis left in a conducting state and the power switching device that isswitched between a conducting and a non-conducting state, can bealternated. For example, the upper power switching device 35a, may beleft in the conducting state for one cycle while the lower powerswitching device 36a is alternately opened and closed, and the followingcycle the lower power switching device 35a, may be left in theconducting state while the upper power switching device 36a isalternately opened and closed. Alternately, freewheeling may beimplemented on other than a per cycle basis. This alternating of thepower switching devices results in reduced switching frequency, extendedservice life of the devices, reduced switching and system losses, andallows the use of slower switching devices.

In this switching scheme the voltage applied across the phase windingvaries from +HVDC to 0 Volts. This type of switching scheme, where thevoltage varies from a given value to zero, is sometimes referred to as"soft-chopping." In the soft-chopping scheme of FIG. 3B(V) where thevoltage across the phase winding varies from +HVDC to 0 the net averagevoltage applied to the phase winding can be controlled to be a desiredpositive, but not a negative, value.

FIG. 3B(VI) illustrates a soft-chopping scheme in which, while currentis flowing in the phase winding, either: (i) the upper power switchingdevice 35a is left in a non-conducting state while the lower powerswitching device 36a is alternately switched from a conducting to anon-conducting state or (ii) the lower power switching device is left ina non-conducting state while the upper switching device 35a isalternately actuated between conducting and non-conducting states. Thealternating of the power switching device that is left in anon-conducting state and the power switching device that is switchedbetween a conducting and a non-conducting state, discussed above,applies equally in this switching scheme with the same resultingbenefits. In this switching scheme the voltage applied across the phasewinding varies from -HVDC to 0 Volts. In the soft-chopping scheme ofFIG. 3B(VI) where the voltage across the phase winding varies from -HVDCto 0 the net average voltage applied to the phase winding can becontrolled to be a desired negative, but not a positive, value.

The particular switching scheme used for actuating the power switchingdevices 35a-c and 36a-c in the power converter of FIG. 3A is determinedby the switching signals applied to the gates of the power switchingdevices by the controller 40. Controller 40 is illustrated in greaterdetail in FIGS. 4A and 4B.

Referring to FIG. 4A, the controller 40 comprises a semiconductorintegrated circuit chip 41 (indicated by the dashed box) and associatedexternal circuitry. In the embodiment of FIG. 4A the semiconductorintegrated circuit chip 41 comprises a digital core 42 that issurrounded by additional circuit elements including comparators 43 and44a-44c. Comparators 43 and 44a-44c are "digital comparators" in thatthe output of the comparators is either a logic level one (when thevoltage at the + input to the comparator is greater than the voltage atthe - input to the comparator) or logic level 0 when the converse istrue.

Control chip 41 may be an integrated circuit chip that includes aproperly programmed microprocessor or microcontroller. In oneembodiment, the control chip 41 consists of an application specificintegrated circuit chip ("ASIC") that is coupled to an externalelectrically erasable programmable read only memory ("EEPROM") thatincludes operational data and instructions from which a controllerwithin the ASIC can operate.

While the embodiment of FIG. 4 illustrates many of the components of thecontroller as located on chip 41 the components of the controller couldbe placed on a number of discrete chips, or analog circuits could beused.

The control chip 41 receives as input the output of the RPT 3 thatindicates the angular position of the rotor. The precise form that theRPT input takes will vary depending on the specific type of RPT used todetect the rotor position. For example, the RPT input could consist of adigital word representing the actual position of the rotor (if a digitalposition encoder is used as an RPT) or simply pulses that indicate thatthe rotor has rotated through a preestablished angular period. In anyevent, the various possible forms of the RPT and the reception of RPTsignals to provide both rotor position and angular speed informationwill be well known to one of ordinary skill in the art, and will not beaddressed herein in detail.

In addition to receiving the RPT signals as an input, the control chip41 also receives via line 2 the externally derived signal representingthe desired speed or torque of the machine. For purposes of discussionit will be assumed that the externally applied signal on line 2 is arepresentation of the desired torque output of the machine althoughthose skilled in the art will appreciate that the control methods andapparatus discussed herein are also applicable to speed control signals.

The externally generated signal representing the desired torque outputof the machine is received by the chip 41 via line 2 and, in FIG. 4A, iscompared to a reference voltage VREF in comparator 43 to produce amodified torque demand signal on line 2'. In the embodiment of FIG. 4Ait is assumed that the externally supplied torque demand signal is apulse width modulated (PWM) signal where the duty cycle of the PWMsignal corresponds to the ratio of the demanded torque to the maximumtorque available from the machine. In this embodiment, the magnitude ofthe V_(REF) voltage signal is set to a level that is between the logiczero and logic one levels that define the PWM signal. Thus, the signalappearing on line 2' is a PWM torque demand signal that has a duty cyclethat corresponds to the duty cycle of the externally supplied PWM torquedemand signal received on line 2.

For certain systems, the externally supplied torque demand signal online 2 will not be a PWM signal but will instead be an analog signalwhose voltage magnitude corresponds to the desired torque output of thesystem. The controller of the present invention is capable of receivingand operating from such signals with little or no modification. Acircuit that would enable the controller 40 of the present invention toreceive an analog signal is illustrated in greater detail in FIG. 4B.

In FIG. 4B only the portion of FIG. 4A relevant to the reception of theexternally supplied torque demand signal is illustrated. In general, thecircuitry of FIG. 4B is the same as that of FIG. 4A except that aresister/capacitor ("RC") filtering network 45 is coupled to theinverter output of comparator 43. For purposes of illustration theinverted output of comparator 43 is illustrated as being supplied by aninverter 46 This inverted signal may be alternately generated in thedigital core 42 of control chip 41 and provided via an external pin.When an analog signal is received on line 2 that is greater than thevoltage at the non-inverted terminal of comparator 43, the invertedoutput of the comparator 43 will be logic high or positive. Thispositive voltage will cause electrical current to flow into the RCnetwork 45, charging the capacitor at a rate defined by the RC timeconstant of network 45. Eventually, the voltage across the capacitor(which is applied to the non-inverting terminal) will exceed the inputanalog voltage of control chip 41 and the output of converter 43 willswitch states, causing the inverted output to go to logic zero (ground)thus draining the charge off of the capacitor. Eventually the voltageacross the capacitor will fall below that of the analog voltage appliedon line 2 and the output of the converter 43 will again switch states.

In one embodiment, the signal on line 2' is applied as a control signalto an up/down counter that is clocked at a given frequency, such thatthe counter counts up when the signal on line 2 is logic high and countsdown when the signal is logic low. In this embodiment, the output at thecounter corresponds to the speed or torque command signal.

Referring again to FIG. 4A, the digital core 42 of the controller 40receives the RPT information and the PWM torque command on line 2 and,based on this information, determine the appropriate firing signals tobe supplied via lines 5a, 5b and 5c to the power switching devices ofpower converter 30. The precise manner in which the control circuitgenerates the firing signals depends on the control mode in which thecontrol chip is operating.

The control chip 41 is capable of operating in several distinct controlmodes including: (i) "L_(MIN) /L_(MAX) Mode"; and (ii) "2/3 Mode," and(iii) "Full Angle Control Mode." The precise mode in which the controlchip 41 is operating determines the type of firing signals that areapplied to the power switching devices in the power converter 30.

In the L_(MIN) /L_(MAX) Mode the energization of the phase windings isaccomplished over an "active cycle" that is defined by predeterminedT_(ON) and T_(OFF) angles. When the L_(MIN) /L_(MAX) Mode is selected,the T_(ON) angle for a given phase winding corresponds to the angularposition of the rotor with respect to the stator when the inductance ofthe phase winding is approximately at its minimum value, L_(MIN). Inthis mode, the T_(OFF) angle for the phase winding corresponds to theangular position of the rotor when the inductance of the phase windingis roughly at its maximum L_(MAX). In one embodiment of the presentinvention each L_(MIN) or L_(MAX) point of the three phase windings isassociated with a specific output signal from RPT 3. In such anembodiment, the controller (when operating in L_(MIN) /L_(MAX) Mode)initiates the active cycle for a given phase winding when the L_(MIN)signal for that phase winding is received and terminates the activecycle when the appropriate L_(MAX) signal is received.

While the T_(ON) and T_(OFF) angles for the phase windings are set andunchanged in the L_(MIN) /L_(MAX) Mode, the peak magnitude of thecurrent that is allowed to flow in each phase winding during its activecycle is allowed to vary as a function of the torque demand signalreceived by the digital core on line 2. Specifically, the digital core42 includes a circuitry for selecting an I_(ref)(pwm) signal thatcorresponds to a PWM desired duty cycle defining a desired peak currentfor a given speed/torque combination. As those skilled in the art willappreciate, the relationship between torque and current in a reluctancemachine is not linear and can vary with the speed of the machine. Assuch, controller 40 includes circuitry that receives the torque demandand speed information and selects an appropriate I_(ref)(pwm).

In one embodiment of controller 40, I_(ref)(pwm) data for various speedand torque points is stored in a sparse matrix stored in memoryaccessible to the controller 40. In this embodiment the speed/torqueinformation is used to address the matrix memory. For speed/torquepoints for which no stored data is available, interpolation routines maybe used to calculate appropriate I_(ref)(pwm) data.

Alternate embodiments are envisioned where the I_(ref)(pwm) duty cyclesignal is calculated from speed and torque data or where curve fittingtechniques are used to define an equation for I_(ref)(pwm) based onmeasured data.

Referring to FIG. 4A, the I_(ref)(pwm) signal generated by the digitalcore is provided via a pin on the control chip 41 to an external RCfiltering network 47 such that an analog signal corresponding to thedesired peak active cycle current Iref.sub.(DC) is generated. Theprecise size and structure of the RC network 47 will vary depending onthe frequency and magnitude of the I_(ref)(pwm) signal on the particularapplication. In one embodiment, the frequency of the I_(ref)(pwm) signalis approximately 40 Khz, and the RC network 47 consists of resistors48a-48d and two capacitors 49 where the values of the resistors are,respectively, 2.2 K Ohms, 100 K Ohms, 20 K Ohms and 20 K Ohms, and wherethe capacitance value of the capacitors 49 is 100 nano-farads.

The single analog signal representing the desired peak current value isprovided via an input pin to comparators 44a, 44b and 44c where it iscompared with the feedback current signals from the three phase windingsA, B and C. The comparators 44a, 44b and 44c provide an indication ofwhether the respective phase currents are greater than or less than thedesired peak current reflected by the Iref.sub.(DC) signal. It should benoted that the use of a single pin to receive a single Iref.sub.(DC)signal that is compared to each of the phase winding currents simplifiesthe construction and reduces the cost of control chip 41 sinceindividual current references are not required for each of the differentphase windings.

As discussed more fully below, in the L_(MIN) /L_(MAX) Mode, the controlchip 41 receives the current comparison signals and controls the firingsignals for the power switching devices such that, when a phase is inits active cycle, the current in the phase winding will be maintained ata level approximately equal to the level corresponding to theIref.sub.(DC) signal. In one embodiment of the present invention, thecurrent in the phase winding during an active cycle is maintained by afixed frequency soft chopping technique wherein the power switchingdevices associated with an active phase winding are both actuated into aconducting state when the current in the phase winding is below thedesired peak level reflected by the Iref.sub.(DC) and one of the powerswitches is rendered non-conductive to allow freewheeling when thecurrent in the active phase winding exceeds the desired peak currentlevel.

The fixed frequency soft-chopping control scheme discussed above is onlyone method in which current control can be accomplished during theactive cycle. Hard-chopping techniques can also be used, although suchhard chopping techniques will, for the same number of "chops", result inapproximately twice the switching losses since two switches must beactivated for each chop. Alternate forms of soft chopping may also beused. For example, the power switching device that is left in aconducting state and the power switching device that is switched betweena conducting and a non-conducting state can be made to alternate toreduce switching system losses, extend the service life of the devices,and allow the use of slower switching devices.

The L_(MIN) /L_(MAX) Mode discussed above is most suitable for operationof the reluctance machine 20 at relatively low speeds where theinductance increases relatively slowly allowing a current to beestablished in the winding relatively quickly. At high speeds where theinductance changes more rapidly over time, application of fixed dutycycle pulses between L_(MIN) and L_(MAX) may result in a current profilein the phase which increases and decreases too slowly, thus torque isnot easily controlled. The L_(MIN) /L_(MAX) mode may be limited for useat relatively low speeds only. As an example, in one application of thecontrol system of the present invention, the L_(MIN) /L_(MAX) Mode isused for rotational speeds up to 375 RPM.

As the rotational speed of the rotor increases, the negative torque thatis generated by the current that remains in an energized phase windingafter the L_(MAX) position can result in inefficient motor performance.As such, as the rotational speed of the rotor begins to increase, it maybe desirable to end the active cycle of a given phase winding prior tothe L_(MAX) position. In the "2/3 Mode", the controller 40 of thepresent invention ensures that this occurs by defining the T_(OFF) anglefor each phase winding to be the rotor position that is 2/3 of theangular distance between the L_(MIN) and L_(MAX) positions. At this 2/3point which occurs prior to L_(MAX), current in the winding can bereduced relatively slowly, while normal forces are low, rather thanquickly driven to zero. In one embodiment of the system of the presentinvention this 2/3 point corresponds to a specific RPT output signal. Inone embodiment, the T_(ON) angle for an active phase in the 2/3 Mode isthe same as that in the L_(MIN) /L_(MAX) Mode (i.e., Lim), but it couldalso be selected independent of the L_(MIN) /L_(MAX) Mode.

While the T_(ON) and T_(OFF) positions are fixed in the 2/3 Mode, thepeak desirable current is allowed to change as a function of the torquedemand signal. The method used by controller 40 for setting the peakdesirable current and indicating whether a given phase winding currentexceeds this value is that same as that previously described above inconnection with the L_(MIN) /L_(MAX) Mode.

While L_(MIN) /L_(MAX) Mode is best suited for low speeds where hightorque output of the system is an important consideration, the 2/3 Modeof operation is best suited for operation where higher speeds areexpected and system efficiency is key. In 2/3 Mode, the angular speed ofthe rotor is greater than the speed where the negative torque producedusing the L_(MIN) /L_(MAX) Mode begins to reduce efficiency of motorperformance, but still sufficiently low that there is enough timebetween the T_(ON) and T_(OFF) positions for a sufficient number ofcurrent chops to be performed such that reliable current control ismaintained. As an example, in one application of the present invention,the 2/3 Mode is used for rotor speeds above 375 RPM but less than 1000RPM. In other applications, the 2/3 mode can be used at very low rotorspeeds in place of the L_(MIN) /L_(MAX) mode for better efficiency andlower RMS phase current.

As the rotational speed of the rotor increases further, a point isreached where the motor inductance and counter emf produced limits theamount of current which can be established in a phase winding usingeither the 2/3 mode or the L_(MIN) /L_(MAX) mode as described above.Further, if a phase winding is de-energized at a T_(OFF) positioncorresponding to either L_(MAX) or 2/3 the rotational distance betweenL_(MIN) and L_(MAX), it is difficult to remove current which may remainin the winding after the point of maximum inductance prior to the nextenergizing cycle of the winding. Chopping may be done at these speeds,but does not provide adequate current control by itself In thissituation, the reluctance machine may be controlled in an "Angle ControlMode" where the energization of the phase winding is defined usingdifferent switching schemes at different rotor positions or angles. Inone embodiment three angles are used to define discrete +HVDC,freewheeling and -HVDC schemes, although other more complicatedembodiments are envisioned, using different angles and more or less thanthree angles. In this "three-angle" example, the three angles thatdefine the energization of a phase winding being controlled in the AngleControl Mode are: (i) the T_(ON) angle that defines the initiation ofthe active cycle; (ii) a freewheeling angle ("FW") where the current inthe phase winding is allowed to freewheel through one of the powerswitching devices and one of the fly-back diodes associated with thewinding; and (iii) the T_(OFF) angle that defines the termination of theactive cycle. Unlike in the L_(MIN) /L_(MAX) and 2/3 Modes, the T_(ON),T_(OFF) (and FW) angles are not fixed and can vary with the speed andtorque demand of the motor in the Angle Control Mode. Moreover, therotor positions associated with the T_(ON), T_(OFF) and FW angles oftendo not correspond to specific RPT signals such that methods must be usedto estimate the rotor's position between successive RPT signal changes.

Because the T_(ON), T_(OFF) and FW angles vary as functions of theangular speed of the rotor and of the torque or speed demand of themachine, the controller must be provided with (or generate) appropriateangle parameters for each combination of rotor speed and torque that canbe encountered while the controller is operating in the Angle ControlMode. One approach to providing such information is to "characterize"the motor by experimentally running the motor in the Angle Control Modeand determining, for several speed/torque points, the appropriateT_(ON), T_(OFF) and FW angles that provide desirable operatingperformance at the various points. That data can then be stored in adigital memory associated with control 41 (e.g., a memory located onchip 41 or an external memory addressable and accessible to control chip41 for use by the digital control core. In one embodiment, the selectedangle parameters are stored in a sparse matrix stored in memoryaccessible to the control chip 41. The controller either operates fromthe angle parameters associated with the speed/torque point most nearthe actual speed/torque point or uses an interpolation algorithm toselect the appropriate angle parameter data for the actual speed/torqueoperating point. Alternate embodiments are envisioned where the T_(ON),T_(OFF) and FW angles are calculated or otherwise derived from the speedand torque information available to controller 40.

The precise method that should be used to characterize a motor foroperation in the Angle Control Mode will vary from motor to motor andcontroller to controller. General techniques for performing thecharacterizing function are known in the art and will not be addressedherein in detail.

Angle Control mode is best suited for high speed machine operation, i.e.when the machine is operating above a predetermined percentage (e.g.,between 1/3 and 1/2) of its maximum operating speed. As an example, inone application of the present invention Angle Control mode is used forcontrol purposes when the angular speed of the rotor exceeds 1000 RPM.

The controller 40 of the present invention may be configured to operatein only one of the operating modes discussed above or to operate indifferent operating modes depending on the rotational speed of themachine. When the latter approach is desired, the controller can besuitably programmed, by storing the speeds at which a mode change is tooccur, in digital memory locations within the controller 40.

As discussed above, abrupt changes in the magnetic flux within thereluctance machine cause unwanted acoustic noise and vibrations. Themajority of these abrupt changes occur when the current in a phasewinding is decreasing from its magnitude in the active cycle to nearzero (the "back-end" of the current waveform), and when the current inan active phase winding is increasing from near zero to its peak valuein the active cycle (the "front-end" of the current waveform). Theincreasing current can cause changes not only in the flux of theenergized winding, but may also cause changes in the flux of theadjacent winding which is energized or a phase which is beingde-energized. The controller 40 of the present invention can control thenet average voltage applied to the current in a phase winding during theactive cycle in the three previously discussed modes such that thechanges in the magnetic flux in an active phase winding or an adjacentphase winding are reduced, which will reduce unwanted machine noise andvibration.

This current control is accomplished at the front-end of the currentwaveform by "ramping-up" the current in a controlled fashion, and at theback-end of the current waveform by "ramping-down" the current in acontrolled fashion. For purposes of discussion, the method and apparatusused by controller 40 to "ramp-down" the current waveform will bediscussed first. The following discussion is applicable to all operatingmodes and is not dependent on the manner in which T_(ON) and T_(OFF) aredefined or selected.

The simplest way to drive down the current in an active phase winding atthe end of the active cycle for that winding is to actuate both powerswitching devices associated with that winding into a non-conductivestate when the rotor reaches the T_(OFF) position for that active cycle.This technique, however, results in the application of approximately thefull -HVDC bus voltage level to the phase winding at T_(OFF) and resultsin a dramatic change in the net average voltage applied to the windingfrom a net positive voltage (whatever the voltage was that was used toestablish the desired active cycle current) to -HVDC. The abrupt voltagechange tends to cause abrupt magnetic flux changes and results inunwanted motor noise and vibration.

Controller 40 is configured to allow the current to "ramp-down" in apredetermined manner such that the current changes in the active phasewinding are not abrupt, but are controlled from the point at which therotor reaches the T_(OFF) angle for the active phase until the currentin that phase is reduced to near zero. In general, this is accomplishedby implementing a switching scheme where, at T_(OFF), a duty cycle iscalculated or captured such that the net average voltage applied to thephase winding just after T_(OFF) is the same as it was at T_(OFF). Theduration of the voltage pulses applied to the winding is thereaftercontrolled such that the net average voltage applied to the windingvaries in a controlled fashion from its positive value at T_(OFF) tozero and then to a negative value until the current in the phase windingis reduced to near zero. The result of controlling the voltage pulsesapplied to the winding is to ramp down the current and the magnetic fluxin a smooth manner, while the result of applying the equivalent netaverage voltage after T_(OFF) is to create slight changes in the currentwaveform from the point when current starts to be removed from thewinding to the point where the current approaches a value near zero.This controlled current change reduces the audible noise and unwantedvibration in the system.

In one embodiment controller 40 uses a soft-chopping scheme to maintainthe current in the phase winding at its desired peak value during theactive operating cycle. In this embodiment, at T_(OFF), controller 40will initiate a hard-chopping scheme, applying voltage pulses to drivethe current down to zero in a controlled fashion. In one embodiment ofthe system described herein, the duty cycle of the hard chopping voltagepulses applied to the winding at T_(OFF) is half of the soft choppingduty cycle at T_(OFF), as a percent of the fixed PWM frequency, plusfifty percent. While the following discussion will be limited to adetailed discussion of this embodiment of controller 40, it will beapparent to those of ordinary skill in the art that the devices andtechniques discussed herein can be used in conjunction with otherchopping or current control schemes.

FIG. 5A illustrates in flowchart form a method 50 in which the controlsystem of the present invention causes the net average voltage appliedto the phase winding to change in a non-abrupt and controlled mannerfrom approximately the net average voltage being applied at the T_(OFF)point to -HVDC. The controller 40 accomplishes this by first determiningan initial duty-cycle of hard-chopping voltage pulses that correspondsto a net average voltage equal to or near the net average voltageapplied at the T_(OFF) point. This initial duty cycle is thereafteradjusted to control the net average voltage applied to the phasewinding, and to "ramp-down" the current in a de-energized phase windingthrough a controlled application of hard-chop voltage pulses.

Prior to the phase winding entering its active cycle, the digital core42 of controller 40 calculates the initial duty cycle of thehard-chopping pulses that will begin at the T_(OFF) point. This isaccomplished in steps 51-53 of FIG. 5A. Since the net average voltagerequired to establish a desired current magnitude in a phase windingvaries with the speed and torque of the machine, for optimum noisereduction, the duty-cycle for which hard-chopping should begin at theT_(OFF) point (which defines the net average voltage at the T_(OFF)point) should vary as a function of both the rotor speed and the machinetorque output. As such, the controller 40 of the present inventioncalculates in step 51 a contribution to the "HARD-RAMP-START" duty cycle(i.e., a contribution to the net average voltage at T_(OFF)) as afunction of the ratio of the output motor torque to the maximum possiblerotor torque. In the specific example of FIG. 5A, the torque-relatedcontribution to the HARD-RAMP-START duty cycle is calculated by aprocessor in the digital core 42 as a linear function of the torqueoutput where HARD-RAMP-START_(TORQUE) CONTRIB. =T_(OUT) /T_(MAX)*M+OFFSET, where M represents a desired slope of a representative linearcurve and OFFSET represents a desired offset. The precise values of Mand OFFSET will vary from machine to machine and may be determinedexperimentally in a manner similar to the "characterizing" processdescribed above where different M and OFFSET values are tested and theoptimal M and OFFSET values for a given machine are determined. M andOFFSET may vary with speed and torque and OFFSET can be zero in someapplications. Characterization techniques may be beneficial since theoptimum relationship between speed, torque, and the desired net averagevoltage at T_(OFF) (defined by HARD-RAMP-START) is often not linear, butis instead defined by higher order polynomial equations.

After calculating the torque contribution to the HARD-RAMP-START dutycycle in step 52, the controller 40 of the present invention calculatesthe speed contribution to the HARD-RAMP-START duty cycle in step 52.This may be done in a manner similar to that previously described abovein connection with the torque contribution to the HARD-RAMP-START dutycycle.

The torque and speed contributions to the HARD-RAMP-START duty cycle aresummed in step 53 to produce the desired HARD-RAMP-START value for thegiven speed and torque. The value of HARD-RAMP-START could alternativelybe calculated using a single function which intrinsically combines bothfactors.

Although the above describes methods for calculating the HARD-RAMP-STARTduty cycle through the use of linear functions, other approaches may beadopted. For example, higher-order equations could be used forcalculation purposes, or HARD-RAMP-START data for various speed andtorque points could be predetermined and stored in a sparse matrix inthe controller 40. As with the angle parameters discussed above,interpolation routines could be used to calculate HARD-RAMP-START valuesfor torque/speed points not pre-stored in the sparse matrix.

While the above examples reflect a HARD-RAMP-START duty cycle thatvaries linearly with the speed and torque of the machine, non-linearrelationships may also be used as well as estimation or adaptive controlmethods. Such alternate schemes, while requiring more computationaloverhead could provide better results. FIG. 5B illustrates severalpossible HARD-RAMP-START duty cycle curves for various speeds andvarious torque output levels. These curves were determinedexperimentally for a given reluctance machine by monitoring the noiseand voltage produced by the machine and selecting the HARD-RAMP-STARTduty cycle that, for a given speed and torque point, produced the leastunwanted noise. Curves such as those illustrated in FIG. 5B could beexperimentally obtained and then stored in a memory accessible to thedigital core 42 of controller 4, or curve-fitting techniques could beused to determine the equation that most closely fits the experimentallydetermined data.

In the controller 40 of the present invention, at the position T_(OFF),hard-chopping at the HARD-RAMP-START duty cycle is initiated, creating aconvex portion or "rounded comer" in the current profile at T_(OFF). Bycreating this rounded corner, rapid changes in the voltage applied tothe winding and the current and fluxes induced thereby are avoided,reducing the rate of change of magnetic flux. As previously discussed,rapid changes in the flux are a primary source of acoustic noise andunwanted vibration. The hard-chopping duty cycle is thereafter adjustedto control the net average voltage applied to the phase winding suchthat it smoothly transitions from the positive value it had at T_(OFF)to a negative value sufficient to drive the current in the phase windingdown to zero.

FIG. 5C illustrates the phase current in a phase winding 55, and thevoltage across the phase winding 56. FIG. 5D illustrates the phasecurrent in a winding 55, and the net average voltage across the phasewinding 58. In 2/3 mode shown, the controlled ramp down creates a smoothtransistion or rounded corner in region 57 at T_(OFF).

FIG. 6A illustrates in flowchart form one method 60 in which thecontrolled-ramp down is accomplished by the controller 40 of the presentinvention. Initially at step 50 the controller determines an appropriateHARD-RAMP-START duty cycle for the actual speed and torque parameters.This may be accomplished by the method 50 discussed in connection withFIG. 5A or any of the otner methods for determining HARD-RAMP-STARTdiscussed above.

After HARD-RAMP-START is selected, the controller then selects in step61 a RAMP-DOWN-GRADIENT parameter that controls the rate of change ofthe duty cycle that is used in the ramp-down. This RAMP-DOWN-GRADIENTparameter controls the rate at which the net average voltage changesfrom positive to negative across the phase winding and, thus, the rateof change of the current in the phase winding and the resulting magneticflux in the machine.

The RAMP-DOWN-GRADIENT parameter may be calculated as a function ofrotor speed and machine torque in a manner similar to that previouslydescribed above in connection with the calculation of theHARD-RAMP-START duty cycle. As before, the slope and offset values ofthe linear equation defining RAMP-DOWN-GRADIENT as a function or rotorspeed and torque can be experimentally determined. The other approachesdiscussed above in connection with the determination of theHARD-RAMP-START value can also be used to determine theRAMP-DOWN-GRADIENT. FIG. 6B illustrates some experimentally derivedvalues for the RAMP-DOWN-GRADIENT for a given reluctance machine, aswell as some additional linear and higher order curve fits. As may benoted the relationship between the optimum RAMP-DOWN-GRADIENT and speedand torque is not linear although a linear approximation may be used forsimplified control purposes. For the example of FIG. 6B, the value ofRAMP-DOWN-GRADIENT is inversely proportional to the rotational speed ofthe machine. In general, the RAMP-DOWN-GRAD should be selected such thatthe negative slope of the current during rampdown increases as the speedof the machine increases. RAMP-DOWN-GRAD may also be selected such thatthe negative slope of the current during ramp-down varies as torqueincreases to improve machine efficiency.

After the RAMP-DOWN-GRADIENT is selected in step 61, the controller 40then determines whether the rotor has reached the T_(OFF) point for theappropriate phase winding. This may be accomplished by comparing adigital value representing the rotor position (that is derived from theRPT signal) with a digital value representing the T_(OFF) point that iseither fixed for L_(MIN) /L_(MAX) and 2/3 Mode or provided to (orcalculated by) the controller for Angle Control Mode. Once thecontroller has determined that the rotor has reached the T_(OFF)position it then sets a PWM-HARD duty cycle parameter to HARD-RAMP-STARTand begins applying power to the appropriate phase winding at aduty-cycle corresponding to PWM-HARD parameter. The controller thenenters into a loop where the controller repeatedly adjusts the PWM-HARDduty cycle to reduce the voltage applied to the phase winding as afunction of the RAMP-DOWN-GRAD in step 65 and returns to step 64 whereit begins to apply voltage to the appropriate phase winding at the newand reduced PWM-HARD. This cycle continues until the PWM-HARD parametercorresponds to a 0% duty cycle such that the full-HVDC voltage isapplied to the phase winding, or until a rotor position is reached where-HVDC is applied.

In one embodiment of controller 40, the HARD-RAMP-START andRAMP-DOWN-GRAD parameters are numbers that are used by various countersand comparators to generate the appropriate firing signals. Circuitrycorresponding to this embodiment is illustrated in FIG. 6C. Thiscircuitry may be contained within the digital core 42 of control chip 41or may be emulated through the use of an appropriately programmedmicroprocessor or microcontroller.

Referring to FIG. 6C, an 8-bit comparator 66 receives at its A input theoutput of an 8-bit up counter 67 and at its B input the output of an8-bit down counter 68. Comparator 66 provides an output signal(HARD-CHOP-FIRING-SIGNAL) that, in the embodiment of FIG. 6B, ispositive (or logic high) whenever the 8-bit value at A is less than Band that is approximately ground (or logic low) otherwise. TheHARD-CHOP-HRING-SIGNAL may then be used by other circuitry (includingdriver circuitry, not shown) to actuate both power switching devicesassociated with the relevant phase winding into a conductive state whenHARD-CHOP-FIRING-SIGNAL is high and to actuate both power devices into anon-conducting state when HARD-CHOP-FIRING-SIGNAL is low.

Up-counter 67 is configured such that the counter counts up from 0 tothe maximum count value and then resets itself at a rate thatcorresponds to the frequency of the CLK signal. Thus, the output ofup-counter 67 and the A input of comparator 66 (referred to asPWM-COUNT), cycles from 0 to a maximum value at a fixed frequency.

Down-counter 68 receives at its data input a digital value correspondingto the HARD-RAMP-START value determined by the controller. Thus, whenenabled by the RAMPDOWN-GRAD as shown below, down-counter 68 will countdown from HARD-RAMP-START to zero at a rate defined by the CLK signal.The enable signal to down-counter 68 is provided by the count equalszero output of another 8-bit down counter 69. Down-counter 69 isconfigured to repeatedly count down from its data in value (which is theRAMP-DOWN-GRAD) to zero. When the counter reaches zero, it generates aC0 pulse that is applied to the enable input of down counter 68 and tothe load input of down counter 69. Thus, down-counter 69 will enabledown-counter 68 for a single clock pulse at a rate inverselyproportional to the value of RAMPDOWN-GRAD. Thus, the lower the value ofRAMPDOWN-GRAD, the more frequently the C0 pulses are generated and thefaster the down-counter 68 counts down. The faster down-counter 68counts down, the faster the duty cycle of the hard-chopping pulses dropsand the faster the net average voltage applied to the phase windingchanges from positive to negative. It should be noted that the circuitryof FIG. 6C would be repeated for each phase winding.

The ramp-down techniques discussed above represent examples of the typesof rampdown techniques that may be used with controller 40.Specifically, the rampdown techniques described above were in connectionwith a hard-chopping scheme that was initiated at T_(OFF). It ispossible to implement a similar controlled rampdown of the phase currentthat began using a soft-chopping scheme at T_(OFF) according to FIG.3B(V) which results in the same net average voltage being applied at theT_(OFF) point as would be provided with a hard-chopping scheme choppingat the HARD-RAMP-START duty cycle, and then ramping down the softchopping duty cycle using techniques similar to those described aboveuntil the net average voltage applied to the appropriate phase windingis zero, or a desired value above zero. The controller could then switchto a hard-chopping scheme to drive the net average voltage negative orcould switch to a soft-chopping scheme, according to FIG. 3B(VI) toaccomplish the same function.

As examples, FIG. 7A illustrates the phase current in a winding 70, andthe voltage applied to the winding 71, where a soft-chopping switchingstrategy is applied at T_(OFF) 72, the duty cycle is ramped to zero, andthen a hard-chopping strategy is employed in region 73.

FIG. 7B illustrates the phase current in a winding 75, and the voltageapplied to the winding 76, where a soft-chopping switching strategywhich applies the +HVDC voltage and a zero voltage is applied at T_(OFF)77, the soft-chopping duty cycle is ramped to zero, and then asoft-chopping switching strategy which applies the -HVDC voltage and azero voltage is employed in region 78.

Different RAMPDOWN-GRADIENT values could be used for the differentsoft-chop and the hard-chop sections. It should be noted that, for agiven hard-chopping duty cycle, the same net average voltage will beproduced by a soft-chopping duty cycle equal to twice the value of thehard-chop duty cycle minus 50% of the full period of a duty cycle. Oneof ordinary skill in the art having the benefit of this disclosureshould be able to implement these alternate ramp down techniques. Theimplementation of the rampdown techniques described above result in alargely convex waveform from the T_(OFF) point, or a "rounded comer,"which has been found to result in reduced noise.

The ramp-down feature of controller 40 is beneficial in reducing noiseand unwanted vibration for all control modes at all motor torque andspeed points. At high rotational speeds, however, it may be desirable todrive the current in a given phase winding to zero as rapidly aspossible following the rotor reaching the T_(OFF) angle for that phasewinding to avoid producing negative torque. As speed increases,RAMP-DOWN-GRADIENT may be reduced to zero, effectively disabling theramp-down functionality at high speeds and allowing the controller torender both power switching devices associated with the phase windingnon-conductive at T_(OFF), thus applying a full -HVDC voltage across thewinding and driving the phase winding current to zero as rapidly aspossible. In one embodiment of the system of the present invention, theramp-down circuitry is disabled at rotor speeds above 1450 RPM.

In addition to providing for a controlled ramp-down of the currentwaveform, controller 40 also allows for a controlled ramp-up of thephase winding current and efficient fixed frequency chopping control ofthe phase winding during the active cycle once the peak desired currentis reached. The basic method used by controller 40 for ramp-up and peakcurrent control is illustrated in flowchart form in FIG. 8A.

Referring to FIG. 8A, prior to the initiation of an active cycle, thecontroller determines a maximum chopping pulse duration for the activecycle that defmes the maximum width of a chopping pulse during theactive cycle as a function of the signal on line 2'. In one embodiment,this parameter corresponds to a soft-chop pulse width that applies thesame net average voltage, or a slightly greater net average voltage, asthe duty selected for the HARD-RAMP-START duty cycle previouslydiscussed. In such an embodiment this PWM-SOFT-MAX duty cycle parameteris calculated to be equal to twice the value of the HARD-RAMP-START dutycycle minus 50% of a fixed frequency period. This PWM-SOFT-MAX value iscalculated in step 81.

The use of a PWM-SOFT-MAX that corresponds to the HARD-RAMP-START isbeneficial because it reduces the relative complexity of the system inthat it does not require extensive calculations to determinePWM-SOFT-MAX. However, in some applications it may be desirable todetermine PWM-SOFT-MAX independently of HARD-RAMP-START using themethods previously discussed for the development of the appropriateHARD-RAMP-START values for different speed and torque combinations.While the following discussion is in the context of a PWM-SOFT-MAX valuethat corresponds to the HARD-RAMP-START value it should be appreciatedthat the PWM-SOFT-MAX value could be separately determined, and that theHARD-RAMP-START value could be determined from the PWM-SOFT-MAX value.

After determining the PWM-SOFT-MAX in step 81, the controllerdetermines, for each phase winding, whether the T_(ON) position for thatphase winding has been reached. When the position information derivedfrom the RPT 3 indicates that T_(ON) for the phase wmding has beenreached, the controller 40 generates firing signals to actuate both ofthe power switching devices associated with the phase winding into aconductive state thus applying the full +HVDC bus voltage to therelevant phase winding at the frequency and duty cycle corresponding toPWM-SOFT-MAX.

As set forth more fully below, fixed frequency chopping is implementedthrough the use of a counter that begins counting and reaches itsmaximum value at the end of the pulse cycle only to begin countingagain. The output of this counter, referred to a CYCLE-COUNT is used inthe controller 40 of the present invention to provide stable,fixed-frequency pulse width modulation.

Once T_(ON) is reached and both power switching devices are renderedconductive, the full +HVDC power will be applied to the relevant phasewinding until one of two events occurs. The controller continuouslymonitors the output of the current comparator 44 for each phase winding(See FIG. 4A) to determine whether the current in the relevant phasewinding has exceeded the I_(ref)(DC) that defines the maximum desirablepeak phase winding current. If the phase current has exceeded thedesired maximum value, determined in step 84, the controller willactuate one of the switching devices into a non-conductive state, step85, and allow the current to decrease slowly as it freewheels throughthe conductive power device and the appropriate fly-back diode. Thepower switching devices will remain in this freewheeling state until thecounter that defines the CYCLE-COUNT counter is reset in step 86. Uponreset of the counter, both power switching devices will again beactuated into a conductive state, step 83, and the control cycle willrepeat.

If the phase current is below the maximum desired value, both powerswitching devices will remain conductive until the CYCLE-COUNT exceedsthe PWM-SOFT-MAX count. When the CYCLE-COUNT is determined to exceed thePWM-SOFT-MAX, step 87, the-power switching devices are actuated into afreewheeling arrangement, step 85, and remain in that arrangement untilthe CYCLE-COUNT counter is reset.

In controller 40 of the present invention, immediately after T_(ON),when the current in the phase winding begins to increase from zero, thePWM-SOFT-MAX count will be reached before the current in a givenCYCLE-COUNT exceeds the I_(ref)(DC) value. Thus, in this ramp-up region,the PWM-SOFT-MAX serves the function of limiting the rate at which thecurrent in the phase winding increases. Eventually the current in thephase winding will reach a point where, for each CYCLE-COUNT, the phasecurrent exceeds the I_(ref)(DC) value before the PWM-SOFT-MAX count isreached. Moreover, because the inductance increases from or slightlybefore or after the T_(ON) position is reached, the point in theCYCLE-COUNT where the phase winding current exceeds I_(ref)(DC) willshift from a value nearer the start of the PWM cycle (when the phasewinding inductance is low) to a point near or past PWM-SOFT-MAX as thephase winding inductance approaches its maximum value. Thus, inoperation, the width of the soft-chopping pulses that are applied to thephase winding will vary from a duty cycle defined by PWM-SOFT-MAX duringthe ramp-up cycle of the current waveform, to a more narrow duty cyclewhen the phase inductance is low (because of the limiting effect ofI_(ref)(DC)), to a duty cycle that is very nearly PWM-SOFT-MAX at theT_(OFF) point. FIG. 8B illustrates current in a phase winding 88increasing at a constant PWM-SOFT-MAX duty cycle until the currentexceeds I_(ref)(DC).

In addition to providing advantages in controlling the RAMP-UP of thephase current, the controller of the present invention's limiting of themaximum chopping duty cycle during the active cycle is beneficialbecause it allows for a flxed frequency chopping scheme that does notsuffer from current programming instability. As those of ordinary skillin the art will recognize, fixed frequency converters suffer from apotential current programming instability which may result in pulsesbeing missed or skipped, or the duty cycle changing abruptly, such thatfixed frequency chopping is not maintained. This problem is wellrecognized and is discussed, for example, in Shu et al. "Modeling andAnalysis of Switching DC to DC Converters and Constant Frequency CurrentProgram Mode" presented at the Power of Electronics SpecialistConference in 1979. Typically complicated analog slope compensationcircuitry is used to address this problem.

The controller of the present invention solves the current programminginstability problem through the use of the PWM-SOFT-MAX limiter and theselected switching strategy implemented by controller 40. In particular,because the chopping pulses in the active region are limited to aspecific maximum duty cycle, fixed frequency chopping will always occur.Moreover, the use of the particular scheme adopted in controller 40insures that any variances between the desired current and the actualphase current will be mnims over time. This is accomplished at lowspeeds by insuring that the PWM-SOFT-MAX value is set at less than 50%of the fixed frequency duty cycle. At higher speeds and higher currentswhere larger duty cycles are required, instability is avoided throughthe use of soft chopping in the active cycle. This soft chopping solvesthe instability problem because the rate of change of current increasein a soft chopping scheme is greater than the rate of change for currentdecrease during freewheeling. This difference in the slopes of thecurrent increase and current decreasing portions solves the instabilityproblems.

FIG. 8C illustrates exemplary circuitry for implementing both theRAMP-UP and RAMP-DOWN control functions discussed above. In general,FIG. 8C is similar to the circuitry of FIG. 6C with the primaryexception being the addition of a multiplexor 91 which receives as itsinput the output of DOWN-COUNTER 68', and a digital signal correspondingto the PWM SOFT-MAX value. A control input 92 determines which of theinputs to multiplexor 91 passes through to the B input of comparator66'. During operation, circuitry not illustrated detects whether thephase winding corresponding to the circuitry of 8C is in the activecycle (defined by the detection of the rotor reaching the T_(ON)position). When the active cycle is detected, the input 92 tomultiplexor 91 is set such that the PWM SOFT-MAX value passes throughmultiplexor 91 to the B input of comparator 66'. Thus, from this pointthe comparator will produce pulses having a width that is defined by thePWM SOFT-MAX value. The output of comparator 66' may be provided tocircuitry, not illustrated, which converts the output of comparator 66'to soft chopping pulses having a duty cycle defined by the lesser of:(i) the point where the phase current reaches the digital valuecorresponding to I_(ref)(DC) ; or (ii) the PWM-SOFT-MAX value. Once theactive cycle ends, as reflected by the detection of the T_(OFF)position, the control input 92 to multiplexor 91 will change such thatthe digital value provided by down counter 68' is passed throughmultiplexor 91 to the B input of comparator 66'. An appropriate enablingsignal may then be applied to DOWN-COUNTER 69' such that DOWN-COUNTER69' is enabled only after the T_(OFF) position is detected. Uponactivation of DOWN-COUNTER 69', the circuitry of FIG. 8C will serve toRAMP-DOWN the phase current in a manner previously described inconnection with FIG. 6C.

While the circuitry of FIG. 8C uses the HARD-RAMP-START value to definethe chopping duty cycle or pulse width at the T_(OFF) point, inalternate embodiments the cycle count at which the chopping pulse justprior to T_(OFF) terminated is detected, and used in place of theHARD-RAMP START value when RAMP-DOWN begins. The detection of the cyclecount at the time the last complete chopping pulse in the active cycleoccurs may be monitored through simple circuitry and loaded asHARD-RAMP-START prior to T_(OFF). The use-of this circuitry, whileslightly more complex, insures that there are no changes in the netaverage voltage applied to the phase winding before and after turn off.

While the above discussion has been generally directed to fixedfrequency hard-chopping and soft-chopping techniques, those skilled inthe art will recognize that other switching strategies could be employedto control the voltage applied to a phase winding. Such switchingstrategies could include fixed on-time or fixed off-time variablefrequency techniques, voltage notching techniques, and the like. Inaddition, alternate embodiments are envisioned, and one skilled in theart will recognize, that current control techniques could be applied inplace of the voltage control strategies employed by the currentembodiment.

FIGS. 9A and 9B illustrates representative current wave forms for amotor operating with and without the RAMP-UP and RAMP-DOWN techniquesdescribed above. FIG. 9A generally illustrates a current wave form wherethe current rises dramatically in the region 93 from a zero value to themaximum peak value. Chopping is then initiated to control the peak phasecurrent until the TURN-OFF point for the phase winding is reached at 94.The power switching devices associated with the phase-winding are thenrendered non-conductive and the current is driven sharply down to zero.As a review of FIG. 9A indicates, there are abrupt current changes atboth the front end and tail ends of the current wave-form which tend toproduce unwanted noise and vibration. FIG. 9B reflects a currentwave-form for which the RAMP-UP and RAMP-DOWN techniques of the presentinvention were applied after the T_(ON) point in the region 93'. Thecurrent does not rapidly increase but instead increases in a controlledmanner to the maximum peak current value. Also, after the T_(OFF) point94' the tail current does not drop off suddenly but is instead rampeddown slowly such that abrupt changes in the applied voltage, the currentin the phase winding, and in the magnetic flux are reduced, resulting ina reduction of unwanted noise and vibration. FIG. 9b also illustratesthe current tail portions 95 corresponding to one RAMP-DOWN-GRADIENTvalue.

The above description and several embodiments of the present inventionare made by way of example and not for purposes of limitation. Manyvariations may be made to the embodiments disclosed herein withoutdeparting from the scope and spirit of the present invention. Forexample, while the above description is directed to a specific switchedreluctance motor system and control device, the present invention isapplicable to any form of reluctance machine regardless of the number ofpoles, pole shape and general layout and to machine systems that includecontrollers constructed through discrete digital components or analogcircuits. The present invention is intended to be limited only by thescope and spirit of the following claims.

What is claimed is:
 1. A method of enhancing the performance of areluctance machine system, the machine system including a rotor, a phasewinding coupled to a voltage bus by power switching devices, the methodcomprising the steps of:(a) selecting a selected net average voltagelevel as a function of the operating parameters of the reluctancemachine; (b) applying the selected net average voltage to the phasewinding when the rotor reaches a first pre-established angular position;and thereafter (c) monitoring the current in the phase winding and, oncethe phase current exceeds a desired peak current value, controlling thecurrent in the phase winding to be near the desired peak current value;(d) applying the selected net average voltage to the phase winding whenthe rotor reaches a second pre-established angular position; andthereafter; (e) controlling the net average voltage applied to the phasewinding as a function of time such that the net average voltage variesfrom the selected net average value to a pre-established voltage levelin a controlled fashion.
 2. The method of claim 1 wherein thepre-established voltage level is the negative of the voltage bus.
 3. Themethod of claim 1 wherein the net average voltage applied to the phasewinding varies from the selected net average voltage to thepre-established voltage level as a linear function of time.
 4. Themethod of claim 1 where the selected net average voltage level isselected as a function of the speed of the machine.
 5. A method ofcontrolling the energization of a phase winding in a reluctance machine,the machine having a rotor and at least one phase winding, the methodcomprising the steps of.(a) controlling the voltage applied to the phasewinding during a first voltage-control interval that begins at a pointdefined by a first pre-determined angular rotor position and ends whenthe current in the phase winding first exceeds a desired peak currentvalue; and thereafter (b) controlling the current in the phase windingto be near a desired peak current value during a current-controlinterval defined by the point where the current in the phase windingfirst exceeds the desired peak current value and ends when the rotorreaches a second pre-determined angular position; and thereafter (c)controlling the voltage applied to the phase winding during a secondvoltage-controlled interval that begins when the rotor reaches thesecond pre-determined angular position and ends when the net averagevoltage applied to the phase winding reaches a pre-established voltagelevel.
 6. The method of claim 5 wherein the net average voltage appliedto the phase winding is a constant selected net average voltage level.7. The method of claim 6 wherein the net average voltage applied to thephase winding at the initiation of the second voltage-controlledinterval is equal to the selected net average voltage level.
 8. Themethod of claim 7 wherein the phase winding is coupled to a DC bus bypower switching device, wherein the pre-established voltage level is thenegative of the DC bus, and the net average voltage applied to the phasewinding during the second voltage-controlled interval varies linearlyover time from the selected net average voltage level to the negative ofthe DC bus.
 9. A method of enhancing the performance of a reluctancemachine system, the machine system including a rotor, a phase windingcoupled to a voltage bus by power switching devices, the methodcomprising the steps of:controlling the voltage applied to the phasewinding during a first voltage-control interval that begins at a pointdefined by a first pre-determined angular rotor position and ends at apoint defined by a second pre-determined angular rotor position;capturing the net average voltage level of the voltage applied to thephase winding when the rotor reaches the second pre-determined angularrotor position; applying the captured net average voltage to the phasewinding immediately following the rotor reaching the secondpre-determined angular position; and thereafter, controlling the voltageapplied to the phase winding such that the net average voltage variesfrom the captured net average value to a pre-established voltage levelin a controlled fashion.